High order modulation schemes, like PAM-m and m-QAM (m>2), are being used in short reach optical communication links in modern data centers for future 400 Gb/s links to achieve higher spectral efficiency with respect to the basic NRZ modulation scheme. Such high order modulation schemes require optical links with strengthened linearity requirement on the transmitter and receiver to not distort the transmitted signal. FIG. 1 illustrates a typical linear optical receiver 1 used in coherent optical receiver comprising a photo diode (PD) 2, a trans-impedance amplifier (TIA) 3, and a variable gain amplifier (VGA) 4, which form the receiver analog front-end (AFE) 5. The optical receiver also comprises an analog to digital converter (ADC) 6 followed by the digital back end 7. The optical signal is received by the PD 2, which generates electric current proportional to the received signal power. The TIA 3 converts the PD current to voltage, which is amplified by the VGA 4 to the desired signal level for the ADC 6. A fixed known signal amplitude is required at the input of the ADC 6 to make correct analog to digital conversion for the following digital back-end 7. The variable gain amplifier 4 is used in the receiver RF chain to control the amplitude of the output signal from the AFE 5 that is fed to the ADC 6. The output signal amplitude from the VGA 4 is set based on the reference voltage of the ADC 6.
Conventionally, linear optical receivers 1 are implemented with an automatic gain control (AGC) loop 8 to fix the receiver AFE output amplitude for the following ADC 6, as shown in FIG. 2. The AGC loop 8 is a negative feedback loop that comprises a peak detector 9 and an error amplifier 11. The amplitude of the voltage signal output from the VGA 4 is sensed using the peak detector 9 and compared with a reference voltage signal (OA) using the error amplifier 11 that drives a gain control signal (GC) of the VGA 4. For large loop DC gain, the AGC loop 8 settles when the output voltage of the peak detector 9 equals the reference voltage signal (OA), which is considered as a controlling knop for the receiver AFE output signal amplitude.
FIG. 3 illustrates a plot of a gain control signal GC for a typical AGC loop 8 versus the input PD current (IPD) at different reference voltage (OA) settings. For the same reference voltage OA signal, the gain control GC signal increases, while the input PD current decreases to maintain the amplitude of the output voltage signal from the VGA 4 constant. The receiver dynamic range is defined as the input PD current range that is affordable by the receiver 1 for a fixed output amplitude as shown in FIG. 3 (between IMIN and IMAX). Moreover, the gain control GC signal value increases with the reference voltage OA signal for the same input PD current. Thus, there are a set of gain control GC values for the same input PD current dynamic range depending on the reference voltage OA signal.
Signals from advanced modulation schemes have high signal to noise ratio (SNR), which must be preserved in the receiver chain until the digital processing, in order to achieve the required bit error rate requirement. Thus, a highly-linear, low-noise optical receiver AFE 5 alongside a high-resolution ADC 6 are required in order not to degrade the received signal SNR. Furthermore, a wide bandwidth receiver AFE 5 is required to avoid any inter symbol interference introduced in the received signal. Consequently, high baud rate coherent optical receivers 1 are characterized using four main aspects: 1) noise, 2) linearity, 3) bandwidth, and 4) dynamic range.
Optical receivers 1 with low noise and high linearity are required in order not to degrade the received signal SNR. For small input PD current levels, the received signal SNR is dominated by the receiver noise; however, for large input PD current levels, the received signal SNR is determined based on the receiver linearity. Typically, the receiver AFE noise is governed by the front-end TIA 3 while its linearity is dominated by the following VGA 4. Thus, a low noise front-end TIA 3 is required to be implemented with a highly linear VGA 4 in the coherent optical receivers 1.
The dynamic range of linear optical receivers is defined as the ratio between the maximum overall trans-impedance gain of the receiver AFE 5 to the minimum trans-impedance gain, which translates to the minimum and maximum photo diode currents that can be amplified by the receiver AFE 5 for fixed output signal amplitude. In coherent optical links, the received optical power can vary between 10 dBm to −15 dBm, which translates to 5 mA to 150 uA photo diode current assuming 0.5 A/W diode responsivity. Conventionally, fixed TIA gain and VGA are utilized in the optical receiver AFE as shown in FIG. 2. However, using one VGA 4 in optical receivers is not sufficient for achieving 25 dB dynamic range, as a single VGA 4 provides, at most, a 15 dB dynamic range.
FIG. 4 illustrates a conventional way to implement an optical receiver AFE 15 with a high dynamic range. Two VGA stages 14a and 14b are utilized for higher receiver dynamic range. The gain from both VGAs 14a and 14b is controlled with the AGC loop 8, such that the gain control signal (GC) of both are driven with same error amplifier 11 (as in FIG. 3). By utilizing two VGA stages 14a and 14b, the achieved receiver overall dynamic range increases as well as its linearity, which improves the received signal SNR for large signal levels. However, this architecture suffers from two main drawbacks: 1) poor noise performance, and 2) bandwidth limitation.
In this architecture, the whole dynamic range of the receiver is obtained by the VGAs 14a and 14b; however, the front-end TIA 3 is implemented with a fixed trans-impedance gain (fixed feedback resistor 12). In order to increase receiver linearity, the value of the TIA feedback resistor 12 is set based on the maximum affordable PD current that can be amplified without degrading the VGAs linearity performance. Therefore, a TIA 3 with a feedback resistor 12 having a small value is utilized in this architecture to improve the receiver linearity in expense of its noise performance, as the TIA noise is inversely proportional to the value of the resistor 12. Large receiver noise degrades the received signal SNR specifically for small input PD current. Consequently, this architecture suffers from noise-linearity tradeoff in determining front-end TIA gain, which limits the received signal SNR either at high input levels or small levels. Furthermore, increasing the number of amplifying stages in the RF chain increases the receiver power consumption as well as it reduces its overall bandwidth.
An object of the present invention is to overcome the shortcomings of the prior art by providing an analog front end with an automatic gain control loop for controlling the gain from both the VGA and the TIA.